The KD2BD 9600 Baud Modem
By: John A. Magliacane, KD2BD
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The KD2BD 9600 Baud Modem
By: John A. Magliacane, KD2BD |
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This paper was originally published in the February, March, and April 1998 issues of Satellite Times magazine.
The KD2BD 9600 Baud Modem is a low-cost, high-performance 9600 bit per
second FSK (Frequency Shift Keying) modem designed to interface between
a standard packet radio terminal node controller (TNC) and an FM voice
transmitter and receiver. The modem uses commonly available components,
allows full-duplex access to the 9600 baud digital communication satellites,
and is suitable not only for digital satellite communications, but for
terrestrial packet radio communications as well.
The KD2BD 9600 Baud Modem went from conception to reality in just seven
days. It uses some of the time-proven signal processing techniques used
in the 1200 Baud KD2BD Pacsat Modem developed several years ago, and
should be of interest to amateurs wishing to add 9600 baud digital
communication capabilities to their satellite or terrestrial packet
radio stations.
The KD2BD 9600 Baud Pacsat Modem was designed with several important
design goals in mind. First, it was designed to use commonly available
components and not rely on special EPROMS for transmit waveform synthesis
or bit clock detection, thereby allowing easy, inexpensive, and uncomplicated
duplication. Secondly, since it has been shown that even randomized 9600
baud baseband data contains a DC component that is ignored in many other
9600 baud modem designs, DC coupling is used throughout the modulator and
demodulator sections of the KD2BD 9600 Baud Modem for optimum performance.
And lastly, the design is essentially uncomplicated, allowing an understanding
and appreciation of its operation by both veteran OSCAR users and beginners
alike.
Before getting into the details of this particular modem design, it is
helpful to look back to the beginning of amateur packet radio communications
to have an understanding of the events and logic that led to the protocol
standards used today in 9600 baud digital communications. When AX.25 protocol
amateur packet radio communications first began in the early 1980s, early
experimenters used Bell-202 type audio frequency shift keying (AFSK)
telephone modems to pass packet binary data over the air using voice-grade
VHF narrowband FM transceivers. Bell-202 modems were selected not for
technical or performance reasons, but because of their wide availability
at the time on the surplus market. A data rate of 1200 bits per second was
used, which was four times faster than what telephone modems were capable
of at the time, and was many times faster than the fastest amateur radio
radioteletype (RTTY) communications. Although Bell-202 modems were originally
designed for 300 bit per second data communications over telephone lines,
they were found to function satisfactorily for half-duplex radio work at
four times their design speed.
When terminal node controllers made their appearance on the commercial
market, they included internal Bell-202 AFSK modems for communications at
1200 bit per second as standard equipment. This was done for compatibility
with the early standards set by the packet experimenters, and because the
Bell-202 AFSK modem protocol was proven to work satisfactorily for packet
communications. Unfortunately, the trend caught on, and while the
transmission rate of telephone modems soared from 300 bits per second
to 56 kilobits per second, packet radio communications stagnated at just
1200 bits per second, causing many hams to turn away from wireless digital
communications in favor of landline-based (Internet) communications.
While some vendors offered packet radio TNCs with 2400 bit per second
capabilities, few people purchased them for fear of being incompatible
with the rest of the packet community. Before 9600 bps digital satellites
came into play, the only people who got involved with higher data rates
were those enterprising individuals who developed high speed packet radio
"backbone" networks on UHF frequencies. Unfortunately, few end users got
the chance to experience high-speed packet radio communications first
hand, and directly witness the significant improvement that could be
made over a "standard" TNC with an internal Bell-202 modem.
The Bell-202 AFSK modem standard that is still widely used for the majority
of 1200 bit per second VHF-FM packet radio communications uses audio tones
of 1200 Hz and 2200 Hz to represent the binary '1's and '0's of packet
radio's HDLC baseband serial data. The use of an audio frequency shift
keying protocol allows binary data to pass over an AC-coupled voice grade
communications link. The method works, but its not without its problems.
It occupies a lot of bandwidth -- so much so that it is possible to pass
data at 9600 bits per second in slightly less bandwidth if a different
modem and RF Frequency Shift Keying (FSK) modulation are used.
At 9600 bits per second, it is not possible to convert the '1's and '0's
of a serial binary data stream into audio tones for application to an FM
voice transmitter and remain within legal RF bandwidth limitations. The
use of AFSK at 9600 bits per second would also exceed the bandwidth
limitations of voice-grade communications equipment. Instead, 9600 bit
per second baseband data is used to directly frequency modulate the RF
carrier of the transmitter.
When early high-speed packet radio communications experiments were conducted
on VHF frequencies over a decade ago, Steve Goode, K9NG, found that directly
modulating an FM transmitter with transmit data generated by a packet radio
terminal node controller was not a very effective way of transmitting
high-speed data. To begin with, the square waveshape of the TTL-level data
generated by the TNC is rich in harmonics and would occupy a very wide
bandwidth if used to directly modulate an RF carrier. Steve found that
passing the transmit data stream through a multi-section low-pass filter
was an effective way of reducing the bandwidth of the transmitted signal
to the minimum required for effective communications. The smaller bandwidth
also reduces the chances of causing interference to adjacent channel users,
and allows the signal to pass through the IF filters of standard narrowband
FM voice receivers without excessive phase distortion.
Steve also realized that the unsymmetrical waveshape of AX.25 baseband data
carried a significant DC component that needed to pass without distortion
through the RF communications link between transmitter and receiver. In
theory, this is not difficult. In reality, however, the frequency stability
of commonly available narrowband VHF-FM communications equipment was found
to be rarely high enough to pass a DC-referenced digital signal without
significant bias distortion. Something needed to be done to reduce the DC
component level of the signal in such a way that would not increase its
bandwidth.
Steve tackled this problem by passing all transmitted data through a
scrambling or randomization circuit prior to transmission and unscrambling
received data to restore the signal back to normal. The scrambler randomized
the data transmitted data pattern, thereby minimizing the chances of
transmitting long runs of '1's or '0's or repetitious patterns containing
significant low frequency energy.
Randomizing techniques are typically identified by the scrambling polynomial
they synthesize. Steve's circuit was based on a 17-bit maximal length linear
feedback shift register (LFSR). A total of eight different maximal length
randomizing techniques can be employed using a single tap 17-bit shift
register. Steve's design used a scrambling polynomial of 1+x^12+x^17,
whereas old 1200 baud Bell-212A telephone modems used a scrambling
polynomial of 1+x^14+x^17. The technique Steve chose has become the
standard for 9600 bit per second digital communications, and is authorized
by the Federal Communications Commission for amateur use. The technique
produces a pseudo-random data sequence of bits that repeats after 131,071
clock pulses, or every 13.65 seconds at 9600 bits per second. A maximum
of 17 ones or 16 zeros can occur in a row with this method of scrambling.
Figure 1 illustrates the design of the type of data randomizer used in
amateur communications.
Figure 2 shows a block diagram of the KD2BD 9600 Baud Pacsat Modem. The
modem consists of a data modulator and demodulator that act independently
of one another. This mutual exclusion allows the modem to be used for
full-duplex satellite communications as well as half-duplex (simplex)
terrestrial communications.
The modulator portion of the modem is very simple. It takes transmit data
(TXD) and clock signals (TXCLK) from the host TNC, applies them to a 17-bit
linear feedback shift register scrambler, and filters the resultant through
a 6th order low-pass filter. This technique produces an exceptionally clean
raised cosine waveshape that is suitable for direct connection to the
modulator of an FM voice transmitter.
The demodulator portion of the modem is a bit more complicated. The
demodulator connects to the detector of a narrowband FM voice receiver,
and gently filters the received signal through a second order low-pass
filter. The purpose of the filter is remove any residual 455 kHz IF
noise that may be present on the demodulated signal. The filtered signal
is then fed to a pair of precision peak voltage detectors that are used
to measure the maximum positive and negative voltage excursions of the
input waveform. The average of the peak excursions represents the voltage
midpoint of the received waveshape regardless of any DC offset present on
the received signal, and is used as a reference in a data slicer and a
digital automatic frequency control (AFC) circuit. The data slicer is used
to convert the received signal to a square waveshape for later processing
in digital logic circuits. The digital AFC is used to slowly tune the
receiver lower in frequency in compensation for Doppler shift when
receiving signals from satellites in low earth orbit.
The processed signal is then diverted into two different directions.
The first direction takes it through a bit clock regenerator, and the
second takes it through an unscrambler and then out to the host TNC for
packet disassembly.
Bit clock regeneration is performed in a very uncomplicated manner. The
filtered and processed received signal is applied to an edge detector
designed around an exclusive-OR (XOR) gate and an RC delay network. The
edge detector produces a short output pulse each time the waveform of
the received signal passes through its voltage midpoint. The waveform
produced by the edge detector is known as a protoclock, and is in phase
with the clock of the received signal. Unfortunately, the waveform produced
is non-continuous, so it is processed through a phase locked loop (PLL)
circuit. The phase locked loop operates at the bit rate of the incoming
signal (9600 Hz), and rapidly locks its oscillator to the average phase
of the protoclock, thereby generating a continuous, noise-free clock
suitable for further processing of the received signal.
The received signal is also passed through a fifth-order low-pass filter.
The filter serves two purposes. First, it attenuates any remaining noise
present on the received signal, and secondly, it delays the incoming
signal long enough so that the center of each received bit is concurrent
with the rising edge of the regenerated clock waveform. This delay is
necessary because the unscrambler is triggered on the rising edge of
the regenerated clock signal rather than the center of each clock pulse.
The filtered signal is then applied along with the regenerated clock to
the unscrambler, the output of which is converted to TTL levels and made
available to the host TNC for final processing.
The low-pass filters used to remove noise from the received signal and
properly shape the waveform of the transmitted signal play a critical
role in the overall performance of the KD2BD 9600 Baud Modem. Considerable
time and effort was spent in testing and optimizing these filters to
adequately process the received and transmitted waveforms without introducing
undesirable products, such as phase distortion. The filters need to have a
flat frequency response up to their cutoff frequency, and must also exhibit
a flat group delay characteristic. Uneven frequency response causes an
unfair bias either toward or away certain bit pattern sequences, while
uneven group delay causes the filtered waveform to undergo uneven
propagation time depending on the frequency of the signal being passed
through the filter. In terms of data communications, uneven group delay
produces phases jitter which causes bit zero crossing point instability,
making receiver bit clock detection and extraction difficult and less
reliable.
Figure 3 shows the frequency response and group delay characteristics of
the receive filter used in the G3RUH 9600 Baud Modem (Issue 3) as simulated
using MicroCAP IV circuit analysis software. Notice the uneven low frequency
response and the excessive group delay distortion below 400 Hz. While the
purpose of the receive filter is to gently remove IF noise from the received
signal, the analysis suggests that the G3RUH receive filter may actually do
more harm than good to the incoming signal. On the transmit side, the G3RUH
Modem uses a Transversal or Finite Impulse Response (FIR) filter to tailor
the transmit waveshape to compensate for distortions present in the IF
filters typically used in commonly available FM receivers.
Figures 4 and 5 show the simulated frequency response and group delay
characteristics of the receive and transmit filters used in the TAPR/K9NG
9600 Baud Modem designed in August 1985. The TAPR/K9NG filters offer a
much flatter frequency response, but the receive filter is still a bit
narrow, and suffers from some low frequency roll-off and group delay
distortion.
Figures 6 and 7 show the simulated frequency response and group delay
characteristics of the receive and transmit filters used in the KD2BD
9600 Baud Modem. The extremely flat frequency response and constant group
delay characteristics of these filters produce an exceptionally clean
raised-cosine transmit waveform and introduce virtually no distortion
to the received signal. These filters were actually designed by empirically
selecting component values that produced the cleanest waveform patterns as
viewed on an oscilloscope. It was only after the filters were designed and
the modem constructed that the computer simulations were run, effectively
confirming their performance and characteristics.
9600 baud Frequency Shift Keying (FSK) modems are typically used in
conjunction with communications quality FM receivers and transmitters.
In actuality, 9600 baud modems are signal processors more than they are
modems (Modulator/Demodulators) since the actual signal modulation and
demodulation processes occur in the groundstation radio equipment rather
than in the electronics that is typically referred to as a "modem".
One of the radical features of the KD2BD 9600 Baud Modem design is that
it uses full DC coupling to the receiver's detector and the transmitter's
modulator. This direct coupling provides a frequency response that extends
for the full spectrum of the FSK signal. DC coupling to an FM receiver's
discriminator and transmitter's varactor diode requires some careful
design considerations. First, let's examine how 9600 baud FSK signals
are received and demodulated.
Demodulators used in receivers for FM detection exhibit an "S Shaped"
response curve. Figure 8 shows the response of a typical FM demodulator.
The demodulator's linear curve extends for several kilohertz above and
below the center of the receiver's final IF frequency. As can be seen
in the figure, an unmodulated carrier applied to an FM demodulator at
its center frequency will produce an output voltage of exactly zero
volts. A carrier either above or below the center frequency will
produce either a positive or negative output voltage. Modulation applied
to the frequency of the carrier produces an AC output voltage from the
demodulator.
Provided the modulation waveform is symmetrical and the carrier frequency
is properly centered, the AC output voltage from the discriminator is
centered about the zero volt level. If the received carrier is tuned to
one side of the response curve rather than the center, the audio output
from the detector will not be centered about the zero level, but instead
be centered about a small positive or negative voltage. This voltage is
known as an offset voltage, and in some receivers is used for automatic
frequency control (AFC) purposes or to drive a zero-center tuning meter.
Modern FM communications receivers typically use a quadrature detector chip
for FM signal demodulation. Quadrature detectors posses the same "S Shaped"
response curve as do classic discriminators, but because they are designed
using active components powered by a single-ended power supply, quadrature
detector output voltages are typically centered about a voltage level that
is several volts above ground. A positive DC offset voltage is always
present, even with properly tuned signals, and this voltage and varies
linearly with receiver tuning.
Even some older transceivers such as the Yaesu FT-726R that use a ceramic
discriminator for FM detection elevate the reference of the discriminator
several volts above ground so that the output voltage is always positive.
With this in mind, we are pretty well guaranteed that the detector output
of an FM receiver tuned to an FSK data transmission will consist of an FSK
waveform riding on top of a positive DC offset voltage (FSK+DC).
Since the position of the FSK carrier within the passband of the FM detector
will vary the offset voltage, the offset will continuously vary over time
when receiving transmissions from communication satellites due to the effects
of doppler shift. Therefore, if a frequency response down to DC is required
and direct coupling is used between the FM receiver detector and the data
demodulator portion of an FSK modem, adequate measures must be taken to
properly handle the DC offset voltage present on the output of the detector
so that it does not interfere with post detection signal processing of the
received FSK data.
The first schematic shows the circuitry used to initially process FSK
signals received by a narrowband FM receiver. Received signals consisting
of a demodulated FSK signal plus a DC offset voltage (FSK+DC) are first
processed through a second order low-pass filter designed around operational
amplifier UlA. The purpose of the filter is to attenuate any intermediate
frequency (IF) noise that may be present on the output of the receiver's
FM detector. A small voltage gain is also provided by this filter, and
a positive DC offset voltage from the host FM receiver is required for
its proper operation.
The output of the low-pass filter is then applied to two precision peak
detectors and a data slicer. The purpose of this circuitry is to convert
the sinusoidal FSK waveform delivered by the receiver to a binary waveform
compatible with digital logic circuitry without being affected by the DC
offset voltage riding along with the FSK waveform.
A positive peak detector is formed around U1B, D1, and U2B, while a negative
peak detector is formed around U1C, D2, and U1D. In operation, the positive
peak detector provides an output voltage equal to the maximum voltage
excursion of the filtered FSK voltage waveform, while the negative peak
detector provides an output voltage equal to the minimum voltage excursion.
Potentiometer R7 allows the average of the two peak voltages ((positive
peak + negative peak) / 2), which is a voltage level equal to the exact
center of the filtered FSK voltage waveform, to be accurately set and used
as a threshold voltage for the data slicer. The beauty of this arrangement
is that if the FSK's DC offset voltage should rise or fall due to receiver
mistuning or the effects of doppler shift, the average voltage prcoduced
by the peak averaging circuit will rise or fall accordingly and always
remain exactly at a level equal to the precise midpoint of the filtered
FSK voltage waveform, thereby insuring proper operation of the data slicer.
Essentially, what this circuitry does is subtract the DC offset voltage
from the incoming signal (FSK+DC), and yield a clean FSK signal that can
then be further processed by the modem.
The data slicer is simply a voltage comparator designed around operational
amplifier U2A that provides a binary output voltage that is a function of
the peak FSK input voltage. The comparator's output voltage goes high if
the received FSK peak positive voltage is above the threshold voltage, or
goes low if the received FSK peak negative is below the threshold voltage.
The output of the data slicer is then fed through exclusive-OR (XOR) gate
U3C which acts as a buffer to produce a high amplitude 12-volt peak-to-peak
voltage waveform with sharp rise times and fall times. The waveform at this
point is essentially a level-converted binary representation of the received
FSK transmission, and could be applied directly to a packet radio terminal
node controller for decoding if it were not for the fact that the 9600 baud
data received was randomized prior to transmission and must first be
unscrambled before processing can take place by the host TNC.
Linear feedback shift register (LFSR) unscrambling circuitry must be driven
by clock pulses that match the frequency and phase of the clock in the
transmitting TNC. Since a 9600 baud clock signal is not directly available
from the FSK transmitter, it must be regenerated locally from the received
FSK data itself using additional circuitry in the modem.
In order to accomplish this, level converted FSK data available from U3C is
applied to an edge detector consisting of an exclusive-OR gate (U6C) and an
RC timing network. The edge detector produces an output pulse for every FSK
logic level transition received. This waveform is known as a protoclock,
and is in phase with the transmitting station's bit clock. It is, however,
non-continuous, and so is used to drive a phase locked loop (PLL) to produce
a continuous clock signal. The phase locked loop is designed around a
micropower 4046B CMOS PLL (U8), and operates at 9600 Hz. The regenerated
clock, which is available on pin 4 of U8, is used to drive the LFSR
unscrambler as well as a sample and hold circuit that is part of a
unique FSK data carrier detection (DCD) circuit.
The center frequency of the phase locked loop's VCO is set to 9600 Hz by
R23. Resistors R24, R25, and capacitors C13 and C14 form a loop filter
that allows the phase lock loop to acquire and maintain lock with the
received FSK signal despite small amounts of noise that may be present
with the incoming signal.
The level converted FSK data available from U3C is further processed
through a fifth-order low-pass filter to match the modem's detector
bandwidth to that of the transmitted signal. The filter also delays
the FSK waveform in time so that by the time the signal reaches the
output of the filter, the bit centers of the filtered FSK waveform
are synchronous with the rising edges of the regenerated clock pulses
and can be sampled at the optimum moment by D-Latch U5B prior to final
processing by the unscrambler. The filter is designed around transistor
Q1, operational amplifiers U2C and U2D, and their associated RC passive
components.
The output of the filter is available on pin 14 of U2D, and if the waveform
present at this point were viewed on an oscilloscope whose horizontal sweep
were triggered on the regenerated FSK clock waveform, an "eye diagram" would
be observed. The "eye," or filtered bit center, is widest at the modem's
sampling point (see Figure 9). The filtered waveform is converted back
to a digital logic compatible square waveshape through comparator U4C, and
applied to an unscrambler consisting of a 4013B D-Latch (U5), a 4006B shift
register, and three exclusive-OR gates. The output of the unscrambler is
converted to 0 to 5 volt TTL levels by Q2, and is made available for final
processing by the host TNC.
Filtered FSK data is also processed through a sample-and-hold circuit
consisting of operational amplifiers U9A, U9B, and a 40168 CMOS bilateral
switch (U10A). The switch is triggered for a very brief instant at the
center of the filtered FSK waveform by a differentiation network consisting
of capacitor C15 and R26. 9600 samples of the voltages taken at the
filtered waveform bit centers are taken and stored in capacitor C16 every
second. The voltage present across C16 is buffered through voltage follower
U9B.
The output of the sample-and-hold circuit feeds an LM3914 LED dot/bar display
driver, which forms the basis of the modem's data carrier detection circuitry.
The LM3914 has 10 outputs, each of which represent a discrete voltage level
of the sampled FSK sampled bit center. The presence and quality of an FSK
signal can be determined by examining the filtered bit center. A bit center
that is wide and noise-free is an indication of reception of a valid, high
quality FSK signal. Noise and other distortion narrows the gap within the
opening of the bit center. A signal that is pure noise will display no
opening at all.
The LM3914's four center outputs represent voltages corresponding to the
central region of the filtered FSK bit centers. These outputs are logically
"ORed" together and converted to a ground-referenced voltage through
transistor Q3. The upper three and lower three outputs of the LM3914 are
also combined together through transistor Q4 to produce a reference voltage
against which the center outputs can be compared. The upper and lower
outputs remain fairly constant regardless of whether noise or a clean
FSK signal are being received. The difference between the two transistor
output voltages is an indication of the signal-to-noise ratio of the received
FSK signal. This voltage ratio is used to drive signal quality meter, M1,
as well as a voltage comparator designed around operational amplifier U9C.
The voltage comparator generates an FSK Data Carrier Detect (DCD) output
signal for use by the host TNC. Hysteresis and some low-pass filtering is
used around the comparator to bolster its immunity against noise or false
signals. The DCD output voltage logic is active low to match the requirements
of most terminal node controllers. A DCD connection to the TNC is required
for half-duplex communications, but required for full-duplex satellite
communications. LED D4 provides the user with a visual indication of FSK
data carrier detection.
The slicer threshold voltage available at the center of potentiometer R7
is applied to a second voltage comparator designed around operational
amplifier U4D. This comparator compares the FSK DC offset voltage with
that of a regulated reference voltage adjustable through potentiometer
R8. Since the offset voltage present at R7 drifts slowly with time due
to doppler shift, the comparator output can be used to tune the downlink
receiver lower in frequency to compensate for the effects of doppler shift.
The AFC system used in the KD2BD 9600 Baud Modem is designed to connect
to transceivers that allow UP/DOWN tuning through the front panel microphone
connector. The modem sends a series of pulses generated by operational
amplifier U9D to the groundstation receiver to tune it lower in frequency
during a satellite pass in compensation for doppler shift. Since the
direction the FSK DC offset voltage changes with receiver tuning may
differ between receivers of different makes and models, an AFC polarity
inverter circuit consisting of CMOS switches U10B, U10C, and U10D is
installed between the output of U4D and the gated pulse generator, U9D.
The inverter may be bypassed through switch SW1.
A selection of positive or negative tuning polarities is provided by the
modem as well. Yaesu transceivers require a 5 volt pulse on the frequency
control line to tune the receiver from the microphone connector, while
others require a switch to ground. Potentiometer R8 is adjusted so
that the AFC circuit properly adjusts the downlink receiver when it is
incorrectly tuned, and becomes inactive after proper compensation is
applied by the modem and the receiver is properly tuned. Since doppler
shift causes the signal received from an earth orbiting satellite to
drift lower in frequency and never higher, the automatic frequency
control system used in the KD2BD 9600 Baud Modem tunes the downlink
receiver in one direction only.
LED D3 provides a visual indication of when the automatic frequency
control circuitry in the modem is active. The LED flashes as frequency
corrections are made to the downlink receiver during a satellite pass.
As stated earlier, 9600 baud FSK data is randomized or "scrambled" prior
to transmission. As in the case of the LFSR unscrambler used in the
receive section of this modem, a clock at 9600 Hz is required for proper
operation of the LFSR scrambler circuitry. However, the TX Clock available
from many TNC modem disconnect headers is at 16 times the transmitted data
rate (153,600 Hz). The x16 clock frequency must therefore be divided by
16 to yield a proper signal for operating the scrambler.
U14, a CMOS 4040B binary counter, is used to divide the x16 TX Clock down
to 9,600 Hz. The counter is synchronized to the TX Data waveform through
a differentiation network consisting of capacitor C26 and resistor R59.
The TTL digital logic levels from the TNC are also converted to the 0
to +12 volt CMOS logic levels required by the modem through transistor
Q7, DC blocking capacitor C25, and resistors R57 and R58.
The LFSR scrambler is similar in design and construction to the unscrambler
used in the receive section of this modem. Scrambled transmit data is
available on pin 4 of U3B, and is processed through a six pole low-pass
filter designed around transistor Q8, operational amplifiers U4A, U4B,
and their associated passive RC components. The output of the filter
available on U4B pin 7 is a pure raised cosine voltage waveform. The
waveform is then attenuated and level converted to match and amplitude
and DC bias level required by the varactor diode in the transmitter's
FM modulator to produce a peak carrier deviation of 3.5 kHz.
Since a direct connection is used between the modem the transmitter's
varactor diode, the modem output voltage levels control both the FSK
modulation level as well as the FSK transmitter's center frequency.
Potentiometer R71 adjusts the DC bias voltage level added to the
modem's FSK output waveform and sets the transmitter's center frequency.
R70 sets both the center frequency and the peak FM deviation level of
the FSK transmitter. These adjustments are not mutually exclusive,
so some skill and patience are required to set each one properly.
The KD2BD 9600 Baud Modem was constructed on a single perforated circuit
board measuring 4.5" x 4.5" inches (11.5 x 11.5 cm) using point-to-point
wiring between individual component leads. Perforated circuit board
construction is a simple, yet effective method of prototyping electronic
circuit designs, and takes only slightly greater construction time compared
to building a kit from a printed circuit board design.
Integrated circuit placement is shown in Figure 10. This layout is by no
means sacred, but is included as a guideline for construction. All dual
in-line package (DIP) integrated circuits should be installed in low-profile
sockets to ease construction and allow initial circuit testing to be
performed without the risk of damaging any of the ICs used in the modem.
The two low-power voltage regulator chips and all transistors may be
safely wired in without the use of sockets.
The KD2BD 9600 Baud Modem was designed to interface with an MFJ model
1270B terminal node controller (TNC). The 1270B is a TAPR TNC-2 clone,
and as such, uses a standard modem disconnect header arrangement to
interface with external modems. Figure 11 shows how connections are
made between the KD2BD 9600 Baud Modem and the MFJ-1270B or other TNC-2
clone. Users of TNCs by other manufacturers should consult technical
literature pertaining to their specific TNC to determine the proper
modem disconnect header connections that need to be made between the
TNC and an external modem such as the one described here.
After all circuit components and IC sockets have been wired on the circuit
board, alignment and testing of the modem can begin. Before installing any
integrated circuits into their sockets, verify that connections made for
+Vcc and GND are correct for each chip. Table 2 indicates the DC power
connections for each integrated circuit in the modem, and may be used
as a reference. This test may be made using an ohmmeter. Also verify
that a short does not exist between +Vcc and GND.
If all connections check out correctly, apply 12 volts DC to the modem
and verify that the output voltage of each voltage regulator is correct
and that the proper +Vcc voltage is being fed to each chip socket using
a voltmeter. Table 2 may again be used as a reference. U8 is the only
chip that is supplied with +8 volts. All others are supplied with 12.
If all is found to be correct, power can be removed, and all chips may
be carefully inserted into their proper sockets.
Connect the modem to the host TNC and connect power to the modem. Adjust
both R70, the FSK deviation control, and R71, the FSK Center Frequency
adjust control, to mid position. An oscilloscope attached to the FSK Output
connection of the modem should display a pseudo-random serial data stream.
If a high impedance audio amplifier and speaker or audio signal tracer is
placed at this point, a hissing sound should be heard if the modulator is
working properly.
Audio loop back testing can be accomplished by connecting the modem's FSK
Output to the modem's FSK Input connection. The permits alignment and
testing of the demodulator using the modulator section of the modem as a
test signal source. With a high impedance voltmeter, measure the DC voltage
at U1A pin 1 with respect to ground. Adjust the Slicer Level control, R7,
until the voltage seen at U2A pin 2 is the same as what is seen at U1A pin 1.
Temporarily remove the audio loop back connection, and connect a frequency
counter to pin 3 of U8. Adjust R23 until the counter reads 9600 Hz.
Re-connect the audio loop back connection. At this point, the Data Carrier
Detect (DCD) LED should be lit and the Signal Quality meter should show
significant up scale deflection.
With a dumb terminal or computer running a terminal emulation software
package connected to the TNC, set the TNC's full-duplex parameter ON
(FULLDUP ON). Proper operation of the modem may be had by trying to
establish a connection with the callsign assigned to your TNC. The
throughput experienced should be flawless.
Proper interfacing between any 9600 baud FSK modem and transceiver will
vary depending on the transceiver circuit design and the make and model
of the transceiver. Mike Curtis, WD6EHR, authored a "9600 Baud Packet
Handbook" that describes 9600 baud packet communications, including modem
connection points as well as modifications to popular transceivers. Mike's
handbook was widely distributed electronically via packet radio BBSs several
years ago, and is probably available today via the Internet. Many Internet
sites contain 9600 baud circuit modifications and connection points for 9600
baud modems for many different types of transceivers. The proper transceiver
interfacing methods or transceiver circuit modifications are left to the
expertise of the reader, but the following general information should be
of use regardless of the specifics of the radio transceiver used in
conjunction with this modem.
FSK signals are tapped off from the receiver's FM detector prior to any
de-emphasis circuits or DC blocking capacitors. The input impedance of
the KD2BD 9600 Baud Modem is high, so the modem should not load down
the FM detector and cause distortion or lack of sensitivity when receiving
FM voice signals while remaining connected to the transceiver as is the
case with some other modem designs.
Transmit data from the modem is injected into the varactor diode associated
with the transmitter's modulator through an RF isolation network. Figure 12
shows a representative isolation network consisting of a 47k ohm resistor
and 100 pf capacitor. The isolation network allows signals from the modem
to control the capacitance of the varactor diode while preventing RF that
appears across the diode from being fed back into the modem.
True frequency modulation is produced when the varactor diode used as
the FM modulator is connected in series or parallel with a crystal whose
associated oscillator directly affects the transmitter's operating
frequency. Transmitters that modulate a varactor diode associated with a
phase locked loop (PLL) voltage controlled oscillator (VCO) cannot produce
FM at low modulating frequencies and should not be DC coupled to this or
any other 9600 baud modem. Transmitters that produce phase modulation
will not produce high-quality FSK signals, and should be avoided.
Varactor diodes used for FM modulation are typically supplied a
temperature-compensated DC operating bias voltage in addition to transmit
audio. The exact method used to do this varies from manufacturer to
manufacturer. The DC output voltage of the modem must match the DC bias
voltage present on the varactor diode so as not to alter its operating
bias or affect the center frequency of the transmitted signal. Potentiometer
R71 adjusts the modem's DC bias output voltage level. This level is also
affected to some extent by R70, the Deviation Level Control. The proper
setting of these controls can be easily made by determining the DC bias
voltage present on the varactor diode and the sensitivity of the FM
modulator.
A 10k linear taper potentiometer placed across a 9 or 12 volt DC voltage
source can be used to vary the bias voltage across a varactor diode and
determine how the changing bias voltage effects the final operating
frequency of the transmitter. The negative end of the battery and
potentiometer combination should be connected to the transceiver's
ground, and the wiper of the pot can be connected to the varactor diode
through an isolation network such as the one shown in Figure 12. Varying
the potentiometer will directly vary the operating frequency of the
transmitter, and a voltmeter placed between the wiper of the pot and
ground will measure the voltage being applied to the varactor.
Using a frequency counter to measure the operating frequency of the
transmitter, first determine and record the DC voltage required for
the transmitter to produce an RF carrier 3.5 kHz below what the transmitter
is set to, and then determine and record the DC voltage required to produce
an RF carrier 3.5 kHz above what the transmitter is set.
Attach a DC-coupled oscilloscope to the output of the modem. With the
modem powered and attached to the host TNC, adjust the Deviation and
FSK Center Frequency controls to produce a waveform with a peak-to-peak
voltage to equal to the minimum and maximum voltages determined in the
varactor sensitivity tests. Once these controls are properly set, the
modem may be connected to transmitter, and the result will be a properly
centered FSK signal with a peak carrier deviation of +/- 3.5 kHz.
If the modem is to be used for terrestrial packet radio communications,
there is no need to make automatic frequency control (AFC) connections
between the modem and the RF transceiver. The same is true if the modem
is to be used for satellite communications and the frequency of the ground
station receiver is under the control of satellite tracking and Doppler
correcting software. If corrections for Doppler shift are not made, then
the modem's automatic frequency control feature may be used to keep the
ground station receiver properly tuned to the satellite's downlink
transmissions.
Two separate digital AFC signal polarities are provided by the modem.
Please consult the operating manual of the transceiver used in conjunction
with this modem to determine whether a positive pulse or a switch to ground
is necessary to tune the receiver lower in frequency via the microphone
connection, and make the appropriate connections between the modem and
the ground station transceiver.
The AFC Adjust control, R8, may be properly set by tuning to an unmodulated
carrier, and adjusting the control until the voltages on pins 12 and 13 of
U4D are equal, or the voltage on pin 14 just toggles between a voltage
close to +Vcc and a voltage close to ground.
Proper tuning of the ground station transceiver may require manipulation
of switch SW1, depending on the heterodyne scheme of the transceiver used
in conjunction with the modem. Once the proper setting is determined, the
switch needs no further adjustment unless the transceiver is changed to
a different make or model.
Switch SW2 needs to be front panel mounted, and is used to turn the
automatic frequency control feature on and off. FSK signals received from
satellites should be first tuned in manually with the AFC turned off. After
the signal is acquired, the AFC can be turned on and the modem will track
the downlink signal for the remainder of the satellite pass. Proper tuning
is indicated when the highest upscale deflection of the Signal Quality
meter is achieved.
Communication with any of the current 9600 baud satellites employing the
FTL0 file transfer protocol (such as UoSAT-OSCAR-22, KITSAT-OSCAR-23,
KITSAT-OSCAR-25, or TMSAT-OSCAR-31) requires that the computer connected
to the TNC used at the ground station run FTL0 client software such as
PB/PG, The Microsat Software Ground station Software Suite (MSGSS), WiSP,
or equivalent software. Pacsat satellite communication software may be
found on the Internet at: ftp.amsat.org. Pacsat ground station software
is not, however, required to communicate with the packet radio bulletin
board carried on-board the FUJI-OSCAR-29 satellite or the Mir space
station.
The cost of commercial radio communication equipment can be quite
prohibitive to prospective amateur satellite operators. While modern
transceivers provide many versatile functions and features, few of these
features are required for digital satellite communication. Considerable
money can be saved by taking a more simplistic approach to amateur satellite
communications rather than solving problems with a checkbook.
Considering the fact that most 9600 baud OSCAR satellites have one or
two uplink channels in the 2-meter FM band, a sophisticated uplink
transmitter is not required to access the 9600 baud satellites. An old
2-meter mobile FM transceiver or a Hamtronics VHF-FM exciter may be modified
to produce FSK along with this modem. Downlink reception is possible using
a modified programmable UHF scanner (along with a low-noise preamplifier).
Ed Krome, K9EK, described an effective and low-cost method of receiving
9600 baud satellite downlink signals using a 70-cm to 10-meter downconverter
feeding a Ramsey Electronics 10-meter FM tunable receiver kit in the
March/April 1992 issue of the AMSAT Journal. The bottom line is that
amateur satellite communications does not need to be expensive. Some
technical savvy and a little "ham ingenuity" can go a long way towards
saving many thousands of hard-earned dollars.
Although not tested, it should be a simple matter of scaling the RC
time constant of the PLL and increasing the cutoff frequency of the
low-pass filters to permit the KD2BD 9600 Baud Modem to operate at
much higher data rates. Of course higher data rates would require
a receiver bandwidth in excess of that available from a narrowband
FM receiver designed for voice communications. However, it isn't
a difficult matter to design and build wideband FM receivers these
days with the wide variety of single chip FM receivers currently
available to experimenters. A simple 10-meter wideband FM receiver
can easily be designed around a Motorola MC3362 chip, for example,
and this can be preceded by a 70-cm downconverter to allow reception
of high-speed data with little difficulty.
Digital OSCAR satellites bring worldwide communications to all corners of
the globe at low cost, and even permit unattended and automated ground
station operation. Many digital OSCAR satellites not only function as
store-and-forward message switches, but also carry earth imaging cameras
and scientific experiments that survey the near-earth environment.
The KD2BD 9600 Baud Modem brings to the Amateur Radio community a new
hardware design capable of providing high performance 9600 baud packet
radio communications that permits access to these exciting digital
satellites for less than the cost of a desk microphone. It is hoped
that the KD2BD 9600 Baud Modem will find good use in amateur radio
stations around the world, and that its design will promote a further
understanding of the digital communication methods used in the Amateur
Radio Service, and foster increased experimentation, development, and
refinement of communications techniques used in both terrestrial and
extra-terrestrial communications.
See you on the birds!
Introduction
Design Goals
9600 Baud Communications
Circuit Overview
Filter Design
Receiving 9600 Baud FSK
FM Detectors
DC Offsets and Level Correction
Bit Clock Detection and Regeneration
Low-Pass Filtering
Data Carrier Detection
Automatic Frequency Control
FSK Generation
Modem Construction
TNC Interfacing
Alignment and Testing
Transceiver Interfacing
Determining Varactor Sensitivity
Modem Operation
Cost Saving Measures
Higher Speeds
Conclusion